Voltage-controlled filter

ABSTRACT

An audio frequency filter having voltage controlled transfer functions providing one or more of low pass, bandpass and high pass characteristics, selective in terms of cut-off frequencies and center frequency, including in cascade a first operational amplifier, a first multiplier responsive to a multiplicative function, a second operational amplifier having a frequency selective transfer function, a second multiplier responsive to said multiplicative function and a third operational amplifier having a frequency selective transfer function, in which negative feedback is provided between the outputs of the second and third operational amplifiers and said first operational amplifier.

United States Patent Uetrecht 1 Nov. 11, 1975 [54] VOLTAGE-CONTROLLED FILTER Stable Design." Electronics (pubv 1. 311711969. pp. 75 Inventor: Dale M. Uetrecht, Cincinnati. Ohio 3 4 Colin. Electrical Design & Musical Applications of l l Asslgneg? Baldwm y Clncmnatlan Unconditionally Stahle Combination Voltage Con- Ohlo trolled Filter1'Resonator. Journal of the Audio Engi- [33 Filed; June 5 1973 neering Societ (pub). 12.11971. Vol. 19. Nov 11. pp.

9Z3927. 1211 Appl. No.1 370.303

lriumr Lam111'nerl\'1iehael J, L \neh [52] US. Cl. 328/167: 84/111; 84/119; Assistant E.\'an1i'aerl. N4 E 84/[)|G H); 338/151; 338/1 0; 330/107 wmm'm' Agent. or Firm-NV. H, Breunig [51] Int. CIR. GlOl-I 1/02: GlUH 5/04. (106G 7/16.

HUBF 1/34 [57] ABSTRACT [58] of 328N58- 167; An audio frequene filter having \oltage controlled 30/107 lU transfer functions pro\ iding one or more of lo\\ pass. bandpass and high pass characteristics. seleeti\e in [56] Referencfis cued terms of cut-off frequencies and eenter frequene in- UNITED STA-[ES PATENTS eluding in easeade a first operational amplifier. a first 3.355.668 11/1967 Bocnscl e111], 33 1 7 multiplier responsive to a multiplieathe function. a 3.569.603 311971 Kern 84.11.19 second operational amplifier having a frequene selee- 3.767.833 1011973 Nohle et al. 84/119 X the transfer function. a second multiplier responsive OTHER PUBLlCATlONS Barnav Operational Amplifiers. pp l3 I. published by John Wiley & Sons. Inc.. 1971. Tow. A StepB v-Step Aeti\eFilter Design." IEEE Spectrum, 11/1969, pp. 64-68. Salerno. Active FilterszPart 7. Analog Blocks Ensure to said multiplieatne function and a third operational amplifier hming a frequent) seleetiie transfer funetion. in which negative feedback is pro\ided between the outputs of the second and third operational amplifiers and said first operational amplifier 12 Claims. 13 Draning Figures 1/161 1 262 i zes 144 l z 141 i 1 9 ll+ g 164 In 142 V Ml m b 3 .155 Vn 7.59 V0 cuzcun' 3/ U.S. Patent Nov. 11, 1975 Sheet 2 of9 3,919,648

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0 FEE UENC D\V\8ERS Y l T 1 I \50 r m NOTE RELEASED \Oms NOTE PLAYED U.S. Patent Nov. 11, 1975 Sheet 8 of9 3,919,648

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US. Patent Nov. 11, 1975 Sheet 9 019 3,919,648

VOLTAGE-CONTROLLED FILTER BACKGROUND OF THE INVENTION It is desirable in electric organs to provide a wide range of tone signal filtering characteristics in response to control voltagesv Particularly. in the present organ system. which includes a single voltage controlled oscillator, the frequency of which depends on which of an array of keys is actuated. it is desired that tone quality be a function of the voltage which controls the oscillator. or of the loudness with which a note is sounded, or of the nature of the attack of a tone. all of which are translatable into voltages. The tone filter characteristics desired include control of cut-off in high or low pass filters, and control of center frequency in a band pass filter, and selectivity or Q.

It is the object of the present invention to provide one filter, inherently simple and therefore economical, capable of achieving all the recited results in response to actuation of selector switches and control of one feedback circuit.

SUMMARY OF THE INVENTION A frequency selective filter having pass band, high pass and low pass transfer functions, at will, on a common input variable dc control voltage, and an adjustable feedback control circuit which controls the O of the filter.

BRIEF DESCRIPTION OF THE DRAWING FIG. 1 is a block diagram of a preferred embodiment of the invention;

FIG. 2 is a circuit diagram of a keyswitch circuitry. a note played detector circuit. and sample and hold circuit included in FIG. I;

FIG. 3 is a circuit diagram of mode selector circuitry included in FIG. I;

FIG. 4 is a circuit diagram ofa wave form shaper included in FIGv 1;

FIG. 5 is a circuit diagram of a voltage controlled oscillator and related circuits of FIG. 1;

FIG. 6 is a partial block. partial circuit diagram of brass filters used in FIG. 1;

FIG. 7 illustrates a waveform derived in FIG. 6;

FIG. 8 is a partial block, partial circuit diagram of the flute filters used in FIG. 1;

FIG. 9 is a modification of one filter of FIG. 8;

FIG. 10 is a circuit diagram of a voltage controlled filter used in FIG. 1;

FIG. 11 is a circuit diagram of a vibrato oscillator, noise source and associated circuitry used in FIG. I; and

FIGS. 12 and 13 are waveforms to assist in describing the oscillator of FIG. 11.

DETAILED DESCRIPTION OF THE DRAWING BLOCK DIAGRAM, FIGURE 1 Reference is made to FIG. I, wherein an electronic organ is illustrated as including a plurality of tone generators I, each of which may be composed of plural independent oscillators, one for each note of the organ, or may involve frequency dividers. In the latter case there are traditionally twelve master oscillators for each generator, covering the uppermost octave of notes, from which lower octave tone signals of a manual are derived by frequency division. In the alternative plifier 5, and a loudspeaker 6, or other acoustic radiating system. The gain of the pre-amplifier I7 is controlled by an expression pedal I8 so that tone amplitude is increased as the pedal is depressed. This much is conventional and is contained in many presently commercial electronic organs.

The present invention includes a tone synthesizer that is activated simultaneously, in superposition. with the conventional organ in response to depression of the same keys on keyboard 2 as the keys that control gating of generators l. The output of the synthesizer is controlled in several operating modes. The various operating modes are selected by means of selector [9. Available modes are: l reiteration; (2) percussion; (3) normal (or continuous); (4) fast attack; (5) slow at tack; and (6) sustain.

Control of the synthesizer tones is in response to activation of keyboard 2 that selects a voltage from voltage divider 3 and applies this voltage to lead 24. The voltage represents, in terms of its magnitude, only the nomenclature of the highest pitch key played on keyboard 2, as disclosed in my copending application, Ser. No. 213,939, filed Dec. 30, I971. The voltage on lead 24 is supplied to sample and hold circuit 8 which provides on its output lead a voltage equal to the voltage on lead 24. The voltage at the output of circuit 8 is derived for a time which endures after all played keys are released. or until a different key combination is struck. The sample and hold circuit 8 is activated in response to control signals provided by a note played detector 7 connected to lead 24 to respond to any change of the highest pitch actuated key. The output frequency of a voltage controlled oscillator (V.C.O.) 9 is established primarily by the control voltage supplied to it by the sample and hold circuit 8.

The V.C.O. 9 provides at its output 10a, a square wave signal, the fundamental frequency of which corresponds with the highest note called for by the actuated keys of keyboard 2. The output of V.C.O. 9 is applied to a frequency divider chain 10, which has several outputs for providing an array of square wave tone signals chordally and octavely related to the signal provided by V.C.O. 9. Typically, and for the purposes of the present disclosure, output frequencies of divider chain 10 correspond with organ footages denotes as 32', 16', 8', 4', as well as the partials 2% and 1 /3.

The tone signals provided by frequency divider chain 10 are applied to two sets of tone color filters, II, I2 and to wave shaper 22. Preset voice filters II for brass (e.g., trumpet, trombone, saxaphone) simulation are responsive to the 32 and 16' outputs of chain 10; filters I2 for flute simulation are responsive to the 16, 8, 4', 2% and l /a outputs of chain 10; and wave shaper 22, for unusual tone simulations. responds to the 32', I6 and 8' outputs. Filters II and 12 and wave shaper 22 include an input circuit for each of the footages sup plied to it. Filters 12 are arranged so that the 16', 8, 4' tones are combined on a first output lead and the 2% and l /a' tones are combined on a second output 3 lead 12/).

Wave shaper 22 is also responsive to signals from noise generator 21 and to key activation pulses from noteplayed detector 7. In addition, the square wave tone signals derived from divider chain are sclectively processed, at the will of the musician, in wave shaper 22 as shortened pulses, or sawtooth waveforms or they may be substantially unmodified. The widths of the pulses are controlled in response to the output of sample and hold circuit 8 so that low note key depressions result in wider pulses than high note key depressions. Wave shaper 22 includes operator activated controls for selecting these various waveforms.

Brightness effects of the brass instruments simulated by filters 11 are achieved with variable waveshaping circuits controlled by depression of expression pedal 18, and for simulation of certain instruments, as a function of time elapse after a key has been played. ln the latter case, as time progresses there is less attenuation of harmonic tones, whereby greater brightness is provided as time progresses after initial key activation. In response to depression of shoe 18, the harmonic content of the tone is increased by another waveshaping circuit so as to provide greater brightness as dynamic level increases. Filters 11 also include means for simulating brass attack characteristics by amplitude modulating the tone signals fed thereto with an envelope that includes plural exponential characteristics.

Filters 12 consist of a series combination of a voltage controlled amplifier and a fixed low pass filter. The gain characteristic of the voltage controlled amplifier is proportional to the output voltage of sample and hold circuit 8. Thus the gain of the voltage controlled amplifier is proportional to the fundamental frequency of the input. This causes the output of the fixed low pass filter to remain constant with regard to fundamental frequency over a given input frequency range while attenuating the harmonics of the fundamental frequency input at a fixed db per octave rate.

Filter 11 and wave shaper 22, as well as linear gates 14 and 15 (which may take a form disclosed in US. Pat. No. 3,549,779) are responsive to control signals derived from note played detector 7, as coupled through mode selector 13, a predetermined time, e.g., milliseconds, after the first note of a key combination has been played. The control signals enable the tone signals to be passed through filter 11 and wave shaper 22, as well as gates 14 and 15; these circuits block passage of the tone signals until derivation of the control signals. Because of the delayed enabling, if several keys are activated substantially simultaneously, e.g., within 20 milliseconds of each other, the tones derived from filter 11, wave shaper 22 and gates 14 and 15 are responsive only to the highest pitch note played, regardless of which key was actually struck first. If the musician selects a continuous, rather than percussive, mode, tonal signals may be derived from gates 14 and 15 and wave shaper 22 as long as a key is depressed or, at the will of the musician, a sustain effect after key release can be provided for tones derived from gate 14 and wave shaper 22. On the other hand, if the percussive mode is selected, control signals supplied by mode selector 19 to gates l4, l5 and wave shaper 22, enable the wave shaper and gate 14 to provide a controllable sustain, while gate 15 provides a fixed, short sustain.

In response to one or more keys being released while one or more other keys remain depressed, the tone signals derived from filter 11 and wave shaper 22, as well as from gates 14 and 15 are shifted to tones corrc sponding with the highest pitch of the remaining depressed keys. ln response to all of the keys being released, the control signals from mode selector 19 are removed from filter ll, wave shaper 2 2, as well as gates l4, l5. Circuitry in sample and hold circuit and note played detector 7 delay the V.C.O. 9 from shifting frequency for a predetermined time, e.g., 40 milliseconds, after release of a key so that if several keys are substantially simultaneously released, tones associated with only the key having the highest pitch are derived, regardless of which key was actually the last to be released. If there is activation of a new key of higher pitch than any other depressed key, tones associated with the new key are derived from filters ll, wave shaper 22 and gates 14, 15, 20 milliseconds after striking the new key even though another key was just previously released.

A random, vibrato tonal effect, on the signal derived from V.C.O. 9 is selectively derived from low frequency modulation oscillator 20 via lead 20a, the center frequency of which can be operator controlled. Random vibrato is particularly effective in simulating the characteristics of certain instruments, particularly brass tones, Modulation oscillator 20 frequency modulates V.C.O. 9 at rate, adjustable from one to 50 Hz. In the random mode, signal from noise generator 21 randomly varies the output frequency of modulator 20 about its center value. The amount of random variation is controllable, with typical maximum deviations of t 15% about the selected center vibrato frequency.

ln the reiterative mode, the output frequency of oscillator 20 is fixed, with no random variations imposed. In response to note played detector deriving a signal to indicate that any note is being depressed, oscillator 20 modulates V.C.O. 9 at a fixed frequency. In synchronism with the fixed frequency modulation supplied by oscillator 20 to V.C.O. 9, reiteration pulses are supplied by oscillator 20 to mode selector [9 which, in turn, under the control of the musician, may enable gates 14 and 15 for reiterative effects while a key is depressed.

Another feature of the circuitry including note played detector 7 and sample and hold circuit 8, is simulation of portamento, i.e., a smooth or continuous transition from one tone to another, in response to keys 2 being played legatissimo. If the keys are played staccatissimo there is no portamentation. The portamento effect is selectively provided by including in the sample and hold circuit 8 an electronically controlled switch that selectively short circuits a charging resistor for a storage capacitor responsive to the note indicating voltage supplied to the sample and hold circuit. If the keys are played legatissimo, note played detector 7 derives a control signal that open circuits the switch, whereby the storage capacitor is charged at a relatively slow rate through the charging resistor to provide a slow transition of the voltage controlling V.C.O. 9. In response to staccatissimo, note played detector 7 derives a control voltage that closes the switch to short circuit the charging resistance. Thereby, the voltage across the storage capacitor changes between voltages in discrete steps and the frequency of the oscillator is accordingly altered.

Another tonal efi'ect provides for automatically flatting a note, i.e., reducing its frequency, as it is initially being voiced, to provide accurate simulation of certain instruments, particularly brasses. The amount of flat ting is directly responsive to the note associated with the depressed key. To these ends, the output voltage of sample and hold circuit 8, indicative of the pitch of the depressed key, is coupled to filters 11, and thence sclectively fed to V.C.O. 9 for a transient period when a new tone is being voiced. During the transient period the depressed key indicating voltage reduces the frequency of the V.C.O. to simulate the fiatting effect.

The output of wave shaper 22 is fed via a voltage controlled filter l3 and volume control circuit 23 to the input of pre-amplifier l7 and thence via power amplifier 5 to loudspeaker 6. The outputs of filters 11 and I2, as coupled through gates 14 and 15, proceed via volume control circuit 16 to preamplifier 17, the gain of which is controlled as a direct function of depression of expression pedal 18 so that as the pedal is depressed, gain and loudness are increased.

Voltage controlled, active filter l3 selectively provides a number of different effects on the tonal output signal of wave shaper 22. Filter 13 includes low pass, band pass, and high pass two pole transfer functions that are provided, either singly or in parallel combina tions, for the signal derived from wave shaper 22. The Q and resonant frequency of all three transfer functions are the same. The Q is preset by the musician, while the resonant frequency may be selectively controlled by any of. the played note indicating voltage derived from sample and hold circuit 8, the gating envelope characteristic supplied to gate 14, or the position of expression pedal 18. The resonant frequency increases as the played note frequency increases, or with increased depression of expression pedal 18, or as the amplitude of the gating envelope increases.

KEY SWITCHES, NOTE PLAYED DETECTOR, AND SAMPLE AND HOLD CIRCUIT, FIGURE 2 Reference is now made to FIG. 2 of the drawing wherein there is illustrated a circuit diagram for the elements included in keyboard 2, voltage divider 3, note played detector 7 and sample and hold circuit 8 of FIG. 1. As in the copending application, the voltage derived on lead 24 is indicative of the highest note selected at a particular time, due to the nomenclature assigned to key switches 31, diodes 32 and the values of resistors 33 in voltage divider 3. Voltage divider 3 is connected to a positive d.c. source at terminal 34. Higher notes are associated with higher voltages.

The note indicating voltage on lead 24 is applied through blocking diode 35 to the base of NPN emitter follower transistor 36. Across emitter load resistor 37 of transistor 36 there is developed a voltage directly proportional to the voltage on lead 24. The voltage across emitter load resistor 37 is fed in parallel to conventional monostable multivibrators 38 and 39 which respectively function as note played and note release detectors.

Monostable multivibrator 38 includes NPN transistors 41 and 42 respectively normally biased to the off and on conditions, respectively. The collector of transistor 42 is connected to the base of transistor 41 via a feedback circuit including capacitor 43 and resistor 44, having values selected such that a positive pulse having a duration of milliseconds is derived in response to a positive going pulse being supplied to the base of transistor 41. The positive going pulse may be derived from the emitter of transistor 36 via the ac. coupling circuit including resistor 45 and capacitor 46 or from 6 monostable multivibrator 39, as fed through the a.c. coupling circuit including resistor 47 and capacitor 48.

The collector of transistor 42 is connected to the base of NPN transistor 49, which is driven into saturation in response to the positive 20 millisecond pulse being derived at the collector of transistor 42. The collector of transistor 49 is connected through resistor 51 to be biased by the dc. voltage at the emitter of transistor 36. Thereby, in response to none of keys 31 being closed, which results in transistor 36 being cut off. a zero emitter voltage oftransistor 36 is fed to the collector of transistor 49, and the voltage at the collector of transistor 49 is maintained substantially at ground level. ln response to any one of keys 31 being closed, the resulting positive voltage on lead 24 causes transistor 36 to conduct sufficiently to cause the emitter voltage thereof to increase to a level sufficient to bias transistor 49 into a state enabling it to be selectively cut off and driven into saturation in response to a pulse from monostable multivibrator 3S. Thereby, in response to none of key switches 31 being closed or in response to a key switch being closed for less than 20 milliseconds, the voltage developed at the collector of transistor 49 is maintained substantially at ground. 20 milliseconds after a key switch 31 closure, the collector voltage of transistor 49 jumps positive in response 'to monostable multivibrator 38 changing state. The voltage developed at the collector of transistor 42 is normally at a relatively low level and jumps to a high level for the 20 milliseconds immediately after closure of a key switch 31; after the 20 millisecond period has elapsed, the voltage at the collector of transistor 48 returns to its low level.

Note release detector 39 includes NPN transistors 55 and 56 which are biased so that transistor 55 is normally in a conducting condition, while transistor 56 is normally cut off. The collector of transistor 56 is connected to the base of transistor 55 via a feedback path including series resistor 57 and capacitor 58, having values selected such that a pulse of approximately 40 milliseconds is derived from collector 56 in response to a negative voltage being applied to the base of transistor 55 by the emitter of transistor 36 via an a.c. coupling network including capacitor 59 and resistor 60, In response to the highest pitch note being released, as indicated by a decreased voltage on lead 24 and at the emitter of transistor 36, a negative pulse is supplied to the base of transistor 55 to drive that transistor into a cutoff condition, whereby transistor 56 is driven to a conducting condition. Transistors 5S and 56 remain in this condition for approximately 40 milliseconds, after which time they return to their normal state. Thereby, for 40 milliseconds after a key is released, positive and negative pulses are respectively derived at the collectors of transistors 55 and 56.

The positive going, trailing edge of the 40 millisecond pulse at the collector of transistor 56 is coupled through capacitor 48 and resistor 47 to the base of transistor 41 to change the state of monostable multivibrator 38. This positive going, trailing edge has the same effect on monostable multivibrator 38 as a positive pulse fed to the monostable multivibrator from emitter resistor 37, causing an additional 20 millisecond delay for a total of 60 milliseconds.

The positive going voltage developed at the collector of transistor 49 is utilized to gate tonal signals from preset voice filters ll, flute filters l2 and waveform shaper 22 into output circuitry. Thereby, there is a delay provided for all voices so that if a number of keys are 7 struck within milliseconds, the tones associated with only the highest pitch key are propagated even though the highest pitched key was not actually first struck. If a number of keys are released within milliseconds of each other, while one or more keys remain activated, the tones for only the highest pitch key still depressed are propagated even though the different keys are released at different times. Also, if all of the keys are substantially simultaneously released, no positive going pulse is derived at the collector of transistor 49 causes a gating of the outputs of filters ll, [2 and 22 such that only the previously voiced key tones are propagated, regardless of the order in which the keys are released.

To control selective d.c. coupling of the note indicating voltage on lead 24 to storage capacitor 71 of sample and hold circuit 8, the voltages at the collectors of transistors 42 and 56 are fed to a flip-flop including NOR gates 72 and 73. Output terminals of NOR gates 72 and 73 are d.c. cross coupled in a conventional manner. One input of NOR gate 72 has a dc. connection to the collector of transistor 42, while one input of NOR gate 73 has a dc. connection to the collector of transistor 55. In response to the collector of transistor 42 changing from a low to a high positive dc voltage, the output of flip-flop 70, derived at the output terminal of NOR gate 73, is driven to a binary one, relatively high voltage state. ln contrast, a positive voltage at the collector of transistor causes flip-flop 70 to be driven so that the flip-flop output has a relatively low binary zero voltage level. Thereby, in response to a key being struck, the 20 millisecond pulse developed at the collector of transistor 42 activates flip-flop 70 so that a positive voltage is derived from the output of NOR gate 73; the positive voltage is maintained until the flip-flop state is altered in response to a positive voltage being derived at the collector of transistor 55, as occurs when a highest pitch key switch 31 is released.

In response to a positive or binary one voltage being derived from the output of gate circuit 73, storage capacitor 71 of sample and hold circuit 8 is connected to be responsive to the note indicating voltage on lead 24. To these ends, the positive voltage derived at the output of NOR gate 73 drives normally cutoff NPN transistor 74 into a conducting state. The collector of transistor 74 is connected to gate electrode 75 of field effect transistor (FET) 76 which functions as a first voltage controlled switch of sample and hold circuit 8. In response to transistor 76 being activated into a conducting state, current is drawn from gate electrode 75 to bias FET 76 into a conducting state.

As disclosed in the previously mentioned copending application, sample and hold circuit 8 includes an input circuit 77 and an output circuit 78, each of which includes three transistors for providing impedance isolation and substantially unity gain. Thereby, capacitor 71 is not loaded by circuit 78.

In response to a new high note key being depressed, the voltage on lead 24 is fed through input circuit 77 and the source drain path of FET 76 to storage capaci tor 71. Thereby, changes in the voltage on lead 24 are coupled to capacitor 71 as long as NOR gate 73 is deriving a voltage indicating that a note is being played.

ln response to a highest pitch note being released, while another note is still being played, capacitor 71 is momentarily decoupled, for 40 milliseconds, from the voltage on lead 24. Momentary decoupling of capacitor 71 occurs in response to transistor 74 being driven into cutoff by the output of NOR gate 73 returning to a binary zero level in response to the 40 millisecond positive pulse derived at the collector of transistor 55. After the 40 millisecond period has elapsed, the state of monostable multivibrator 38 is altered, whereby a positive pulse is derived at the collector of transistor 42. The positive pulse is coupled to the input of NOR gate 72, causing flip-flop to change state back to the binary one condition. In response to the flip-flop 70 being returned to the binary one state, FET 76 again switches to a closed state and capacitor 71 is charged to the voltage of lead 24. Thereby, transient changes in the release of key switches 31, while one note remains depressed, are decoupled from capacitor 71 and the capacitor is responsive only to the voltage on lead 24, 40 milliseconds after the release has been performed.

The voltage on capacitor 71 is maintained fixed at a value corresponding with the highest pitch note after all keys are released because FET 76 is open circuited in response to all keys being released. To these ends, normally cut off NPN transistor 81 has its base connected to lead 24. ln response to any of key switches 31 being closed, transistor 81 is driven into saturation, whereby its collector is substantially grounded. The negative going voltage at the collector of transistor 81 is fed through speed-up capacitor 82 and its shunted resistor 83 to the base of normally conducting NPN transistor 84, the collector of which is connected to shunt the base of transistor 74. In response to any note being played, transistor 84 is cut off, whereby the output voltage of NOR gate 73 controls the conducting state of transistors 74 and therefore FET 76. If none of key switches 31 is closed, transistors 81 and 84 are respectively biased to the off and on states, whereby transistor 84 shunts the emitter base path of transistor 74 to hold transistor 74 in a cutoff concondition and prevent FET 76 from conducting.

In summary, in response to a key switch 31 being closed, F ET 76 is closed, whereby capacitor 71 is charged to the voltage on lead 24, causing voltage controlled oscillator 9 to oscillate at the frequency determined by the voltage on lead 28. In response to a note being released which causes a voltage on lead 24 to decrease, a 40 millisecond delay in a change of the voltage on capacitor 71 occurs, because FET 76 is open circuited for the 40 millisecond period.

If, after the 40 millisecond period has elapsed, no note indicating voltage is derived on lead 24, FET 76 remains open circuited and the voltage across capacitor 71 is maintained at the level corresponding with the previous highest pitch played note. If there is still a note depressed after the 40 millisecond period, the voltage corresponding with the new, lower pitch note is fed through FET 76 to capacitor 71. The voice corresponding with the new note is sounded 20 milliseconds after the voltage corresponding with the note is stored on capacitor 7l, by virtue of the positive going voltage derived at the collector of transistor 49. If during the 40 millisecond delay period associated with note release detector 39, a new note is played which causes the voltage on lead 24 to increase, FET 76 is immediately closed and the voltage across capacitor 71 is driven to the new value. 20 milliseconds after the new, higher pitch note has been played, a positive going pulse is derived at the collector of transistor 49 to enable tones associated with the new note to be derived.

A further feature of sample and hold circuit 8 is simulation of portamento. To these ends, the source drain path of PET 76 is connected to capacitor 71 through variable resistor 91 that is connected across the source drain path of FET 92, which functions as an electronic switch in response to the output voltage of monostable multivibrator 93, included in note played detector 7. In response to legatissimo playing, as detected in a manner described infra, the source drain path of PET 92 is open circuited, whereby capacitor 71 is charged through resistor 91. ln response to staccatissimo playing, the source drain path of FET 92 functions as a short circuit for resistor 91, whereby the voltage of capacitor 71 is changed in discrete steps, substantially instantaneously. In response to instantaneous step changes in the voltage of capacitor 71, the frequency of oscillator 9 is stepped. ln contrast, a slow variation in the change in the frequency of oscillator 9 occurs in response to a smooth transition of the voltage across capacitor 71. The charging rate for capacitor 71 while portamento occurs is selected by the musician adjusting the value of resistor 91 to achieve the desired rate of change in the frequency of oscillator 9.

Detection of the legatissimo playing is provided by connecting the input of monostable multivibrator 93 to the collector of transistor 81 through an ac. coupling circuit comprising capacitor 94 and resistor 95. Monostable multivibrator 93 includes NPN transistors 96 and 97 respectively normally biased to the conducting and non-conducting states. The collector of transistor 97 is connected to the base of transistor 96 by capacitor 98 and resistor 99, having values selected so that monostable multivibrator 93 derives an 8 millisecond pulse in response to each negative transition at the collector of transistor 81. Thereby, each time a new set of keys is depressed while no other keys are depressed, monostable multivibrator 93 is activated to derive an 8 millisecond negative pulse at the collector of transistor 97. The negative pulse is d.c. coupled to the gate 100 of FET 92, causing the FET to be driven into a conducting state, short circuiting resistor 91. Thereby, in response to all keys of one set of keys being released prior to any keys of a second set being depressed (i.e., staccatissimo note playing), capacitor 71 is instantly charged to the voltage associated with the new note on lead 24. If, however, the notes are played so that one key is not released until a second key has been depressed (legatissimo note playing), negative pulses are not derived at the collector of transistor 97 and a high impedance path exists through resistor 91 to capacitor 71 to provide gradual voltage change.

MODE SELECTOR, FIG. 3

Reference is now made to FIG. 3 of the drawing wherein there is illustrated a circuit diagram of a portion of mode selector 19. The circuits illustrated in FIG. 3 are concerned with controlling trigger voltages for linear gates 14 and 15, as well as for a-gate included in wave shaper 22. Switches included in mode selector 19 and which functionally are related to other circuits of the system are not illustrated in FIG. 3, but are illustrated in circuit diagrams associated with the particular circuits.

Mode selector 19 includes a monostable multivibrator comprising NPN transistors 102 and 103. The collector of transistor 103 is connected to the base of transistor 102 through a se ries circuit including resistor 104 and capacitor 105, having values selected so that the monostable derives a short duration pulse, on the order of 8 milliseconds, in response to normally cut off transistor 102 being driven into a conducting state in response to a positive pulse being applied to its base. The resulting, short duration pulse derived at the collector of transistor 103 is coupled to the base of inverting NPN transistor 106, the collector of which is connected to the base of PNP transistor 107 which is normally biased into a cut off condition. Transistor 107 is activated into a conducting state in response to transistor 106 being driven into a conducting state, whereby transistor 107, when turned on, functions as a constant current source. To provide different attack rates the current derived from the collector of transistor 107 is controlled by varying the impedance of the transistor emitter circuit. In a slow attack configuration. the emitter of transistor 107 is connected to a positive +30 volt d.c. source through 4.7K resistor 108, and in a fast attack configuration, while switch 111 is closed, the emitter of transistor 107 is connected to the +30 volt source through the parallel combination of resistor 108 and 1K resistor 109, whereby the current of the second configuration is approximately five times that of the first.

The constant current derived from the collector of transistor 107 is applied to a pair of ramp forming networks 112 and 113 which are connected to trigger inputs of gates 14 and 15, respectively; the ramp derived from network 112 is also applied as an enabling input to the gate of wave shaper 22. Each of networks 112 and 113 includes the parallel combination of a capacitor and a resistor; the capacitor and resistor of network 112 preferably have values of one microfarad and 2.2M. and the capacitor and resistor of network 113 preferably have values of approximately 0.05 microfarads and K. so that the charging and discharging rates of the former are considerably less than the latter. The voltage across network 112 is decoupled or isolated from the voltage developed across circuit 113, by virtue of diode 114, the anode of which is connected to network 1 l3, and the cathode of which is connected to network 112.

To control the decay rate of the ramp voltage derived from circuit 112 and thereby provide a sustain effect, potentiometer 114 is connected to circuit 112 via coupling resistor 11S and isolation diode 1 15av Potentiometer 114 is selectively connected to a +30 volt d.c. source by switch 116. The position of slider 117 of potentiometer 114 controls the decay rate of the trailing edge of the ramp voltage derived by circuit 112.

The conducting state and current magnitude of transistor 107 are determined by the system mode of operation, as is the presence or absence of the sustain effect, as derived across network 112. In response to the system being in a percussive mode, transistor 107 is activated into a conducting state for an 8 millisecond period in response to each positive going transition at the collector of transistor 49, FIG. 2, and switch 111 is closed so that a relatively large current is derived from transistor 107. The relatively large current derived from transistor 107 causes capacitors of circuits 112 and 113 to be quickly charged to a relatively high voltage, to provide fast attack and rapid opening, i.e., enabling, of gates 14, 1S and the gate of waveform shaper 22. These results are achieved by connecting the collector of transistor 49 to the base of normally cut off transistor 102 through an ac coupling network comprising capacitor 118 and resistor 119.

After the 8 millisecond on-time of transistor 107 has elapsed, the enabling voltages supplied by circuit 112 to gate 14 and wave shaper 22 and the enabling voltage supplied by circuit 113 to gate decrease. Because of the relatively small resistor and capacitor circuit 113. the decay of the voltage supplied to gate 15 is relatively fast. to cause relatively rapid cutoff of gate 15. In contrast. the resistor and capacitor of circuit 112 are selected to enable a sustain effect to be provided. In the percussive mode. the sustain duration is variably controlled by adjusting the position of slider 117 and closing switch 116. With the position of slider 117 adjusted toward the top of slider 1 14 a relatively long sustain effect is provided, whereby gate 14 and wave shaper 22 remain activated for an appreciable time period to simulate the sustain effect of a percussive instrument.

In a second mode of operation. the continuous mode. gates 14 and 15 and the gate in wave shaper 22 are enabled milliseconds after any of key switches 31 are depressed and remain enabled until all keys are released. To these ends. the collector of transistor 49 is connected via a dc. path to the base of transistor 102 through resistor 119 and switch 121 which shunts capacitor 118 and is closed when the system is operated in a continuous mode. In response to any of key switches 31 being depressed for more than 20 milliseconds, a positive voltage is derived at the collector of transistor 49 and is coupled through switch 121 to drive transistor 102 into a conducting state. Transistor 102 remains in the conducting state as long as a positive voltage is applied to its base. whereby transistor 107 is biased into a conducting state to supply constant current to circuits 112 and 113; thereby. gates 14, 15 and the gate in waveform shaper 22 pass tone signals supplied to them as long as any of key switches 31 are depressed. In the continuous mode. the attack rate can. at the option of the musician, be either slow or fast by opening or closing switch 1 11. Similarly, by closing and opening switch 116, the sustain effect can be a relatively long controllable time or a fixed shorter time for tones fed through gate 14 and the gate of wave shaper 22.

In the reiterative mode, fast attack enabling voltages are derived from circuits 1 12 and 113 periodically, at a frequency determined by the modulation oscillator 20 (FIG. 1), for as long as the key switches remain activated. To these ends, switch 111 is closed and the base of transistor 102 is ac. coupled by series connected re sistor 119, capacitor 122 and switch 123 to be responsive to square waves derived by modulation oscillator 20. Switch 123 is closed by the musician when the system is in the reiterative mode, at which time modulation oscillator 20 is adjusted to derive a random frequency output, whereby monostable multivibrator 101 derives an 8 millisecond pulse in response to each positive going transition of the modulation oscillator output square wave. Enable voltages are derived by circuits 112 and 113 in response to monostable multivibrator 101 being triggered by square wave input in the same manner as described for activation of the multivibrator in response to the positive going trailing edge of the voltage derived at the collector of transistor 49.

WAVEFORM SHAPER 22, FIG. 4

The circuit of FIG. 4 responds to the tones derived from the 8'. 16' and 32' outputs of frequency divider chain 10 to selectively synthesize sawtooth voltages. pulses and square waves having the same fundamental frequency as the selectively coupled tone input signals The duration of the pulses is dependent upon the note of the highest pitch key. The derived square waves are substantial replicas of the input square waves. The different waveforms derived in response to the square wave tones fed to FIG. 4 enable unusual tonal effects to be attained and different musical instruments to be simulated; for example. the sawtooth. pulse and square waves can be utilized to respectively simulate piano. oboe and clarinet instruments. These waveforms can be selectively modified by volt age controlled filter 13 to provide other unusual effects. Waveform shaper 22 is also selectively responsive to random, i.e., noise signals. as well as a pulse each time the state of monostable multivibrator 38, FIG. 2, is changed in response to a new high note signal level being derived on lead 24. The noise input can simulate acoustic effects of. for ex ample. the wind, surf, or a brush hitting a snare, while the pulse derived from monostable multivibrator 38 can provide click effects or noise.

Control of which one or combination of the 8', 16' or 32 tones from frequency divider chain 10 into waveform shaper 22 is provided by selectively forward biasing diodes 131, 132 and 133, having cathodes respectively connected to the 8', 16' and 32' outputs of fre quency divider chain 10. Diodes 131-133 are selectively forward biased by applying positive voltages to anodes thereof in response to closure of normally open switches 134, 135 and 136, which are respectively connected to the diode anodes via resistors 137 so that a +5 volt d.c. level at terminal 138 can forward bias the diodes. The tone signals fed through diodes 131133 are fed in parallel to wave synthesizing networks 141, 142 and 143, which respectively enable selective derivation of sawtooth, pulse and square waves.

Sawtooth wave generator 141 includes NPN transistors 144 and 145 which are responsive to the square wave inputs fed through diodes 131-133 to derive a pulse having a width indicative of the footage passed by the diodes. To these ends, the anodes of diodes 131, 132 and 133 are connected. via capacitors 146, 147 and 148, to the base of transistor 144 which is normally forward biased by the connection of resistor to the +30 volt d.e. source at terminal 149. The values of cal pacitors 146, 147 and 148 are selected in conjunction with the value of resistor 150 to provide an on-time for pulses derived by transistor 144 such that the pulse width is directly related to the footage of the input tone. i.e.. the pulse width for the 8' tone is narrower than the pulse widths for the 16' and 32' tonesv Transistor 145 is selectively maintained in a saturated, forward biased condition by connecting its base to the +30 volt supply at terminal 149 through resistor 152 and switch 153, which is opened by the musician when he wants to synthesize sawtoothtype waves. When switch 153 is closed the collector of transistor 145 is grounded. whereby the base of transistor 144, which is shunted by the collector of transistor 145, cannot be forward biased and sawtooth variations cannot be derived.

To enable the sawtooth, as well as the other waveforms derived by the waveform shaper 22 to be derived, gating NPN transistor 153 is provided. The base of transistor 153 is connected to be responsive to the voltage developed across network 112, FIG. 3, so that transistor 153 is driven into a conducting state from a normally non-conducting state only while a positive voltage of sufficiently high level is derived from network 112. As described supra the voltage level derived from network 112 controls attack rate and sustain times and is dependent upon the system operating 13 mode. In response to the base of transistor 153 being forward biased, current flows from the +30 volt source connected to the transistor collector, to the transistor emitter and thence to the collector of transistor 144 via resistor 154, to enable current to be delivered to the collector of transistor 144.

In operation, negative going transitions of the square waves coupled to the base of transistor 144 from the anodes of diodes 131-133 drive transistor 144 into a cut off condition. Transistor 144 remains in a cut off condition until the voltage across the capacitor 146, 147 or 148 responsive to the square waves fed through the forward biased one of diodes 131-133 reaches a voltage sufficient to activate transistor 144 into a saturated state. Thereby, the cut off duration is determined by the values of capacitors 146-148 and resistor 1S0, whereby a positive voltage is derived at the collector of transistor 144 for a time period indicative of the footage fed to the base of transistor 144 and the widths of generated pulses are accordingly controlled. The frequency of the generated pulses equals the frequency of the input square waves.

The voltage at the collector of transistor 144 is fed to an integrating circuit including resistor 158 and capacitor 159. In response to the positive voltage being derived at the collector of transistor 144, capacitor 159 is charged and the capacitor is subsequently discharged through the emitter collector path of transistor 144 in response to the transistor returning to a saturated condition. The time duration of the increasing ramp derived by the integrating capacitor 159 is determined by the duration of the pulse derived at the collector of transistor 144, while the decay rate of the sawtooth wave derived from the integrating capacitor is substantially constant. Thereby, the leading edge duration of the sawtooth wave is controlled by which of footage is fed through diodes 131-133, while the sawtooth frequency equals the fundamental frequency of the square wave input from divider chain 10. The sawtooth waveform developed across integrating capacitor 159 is coupled to output terminal 164 by diode 161, the cathode of which is connected to resistor 162, and a relatively large d.c. isolating capacitor 163.

To derive pulses having widths determined by the note of the highest note depressed key, the anodes of diodes 131-133 are connected to NPN transistors 165 and 166 which are interconnected with each other, as well as the tone signal and power supply voltages, in a manner similar to the connections of transistors 144 and 145. In particular, the base of transistor 165 is connected to the anodes of diodes 131, 132 and 133 by capacitors 167, 168 and 169, having values selected with criteria similar to those for determining the values of capacitors 146-148. The charging rate of capacitors 167-169 is controlled by the amplitude of the highest note indicating voltage on lead 24, which is d.c. coupled to the base of transistor 165 via terminal 171 and resistor 172 to control the extent of base forward bias of the transistor. In response to variations in the amplitude of the voltage at terminal 171, the cut off time of transistor 165 is varied. In response to a negative going pulse being supplied through one of diodes 131-133 and capacitors 167-169, to the base of transistor 165, the transistor is driven into cut off and remains cut off until the voltage across one of capacitors 167-169 reaches a level sufficient to cause the transistor to be forward biased and driven into saturation, which occurs at a time controlled by the voltage on terminal 14 171, value of resistor 172 and which of the capacitors is responsive to the square wave input.

To enable transistor to be selectively activated and cut off to provide derivation of the pulses, switch 173 is connected between a positive dc. power supply voltage and the base of transistor 166. In response to switch 173 being closed, the emitter collector path of transistor 166 shunts the emitter base junction of transistor 165 to prevent conduction of transistor 165. Collector current flow of transistor 165 is controlled in response to the enable voltage derived by network 112, in a manner similar to that of transistor 144, by the connection of the collector of transistor 165 to the emitter of transistor 153 through resistor 174. Pulses derived at the collector of transistor 165 are fed to output terminal 164 via a pulse shaping circuit including diode 175 which is connected to load resistor 176, the voltage across which is coupled to the output terminal via a low impedance a.c. coupling circuit comprising capacitor 177 and resistor 178.

The derivation of replicas of the square wave voltages selectively coupled through diodes 131-133 is performed with circuit 143, that includes NPN transistors 179 and 181, connected in a manner similar to transistors 165 and 166. The base of transistor 179 and collector of transistor 181 are connected to the anodes of diodes 131, 132 and 133 by a dc. path including diodes 182 and current limiting resistors 183; the cathodes of diodes 182 are connected to the base of transistor 179 to isolate the base from negative transients that might be derived from capacitors 146-148 and 167-169. Square waves are derived at the collector of transistor 179 by the musician opening switch 184, and are fed through the circuit including diode 175, resistors 176 and 178 and capacitor 177 to terminal 164 in response to current being supplied to the collector of transistor 179 by the emitter of transistor 153.

The noise signal from noise generator 21 is selectively coupled to output terminal 164 under the control of the states of transistor 153 and switch 188. To this end, the output signal of noise generator 21 is fed to the base of NPN transistor 185 via an ac. coupling circuit including capacitor 186 that is connected to the noise source. Transistor 185 is normally forward biased by the connection of its base to the plus dc. power supply through resistor 187. Forward bias for the base emitter junction of transistor 185 is removed by closing switch 188, which causes normally cut off transistor 189 to be forward biased and shunt the emitter base junction of transistor 185, thereby driving transistor 185 to cut off. In response to switch 188, however, being open circuited, the noise input signal is fed to terminal 164 via diode 175, resistors 176 and 178 and capacitor 177 when transistor 153 is conducting, by virtue of the dc. connection between the collector of transistor 185 and the emitter of transistor 153.

To derive a relatively short duration pulse each time a new high pitch key is struck, the base of NPN transistor 191 is ac. coupled via capacitor 192 and resistor 193 to the collector of transistor 42 of note played detecting multivibrator 38, FIG. 2. Transistor 191 is normally biased to a conducting state by the connection of its base to the positive dc. power supply via resistor 194. In response to the trailing, negative going edge of the pulse derived at the collector of transistor 42, which occurs 20 milliseconds after monostable multivibrator 38 is driven into a transient state in response to a new high note being struck, transistor 191 is driven to cut off and a positive pulse is supplied to terminal 164 through diode 175, resistors 176 and 178 and capacitor 177. The duration of the pulse is determined by the values of capacitor 192 and resistor 193, components which control the length of time transistor 19] remains cut off. The pulse can be derived only when transistor 153 has been driven into a conducting state. To prevent the pulses derived in response to each activation of monostable multivibrator 38 being coupled to output terminal 164, the emitter base junction of transistor 191 is selectively shunted. Shunting occurs in response to the emitter collector path of transistor 195 being biased into a conducting state by the musician closing switch 196 that selectively connects the positive dc. power supply voltage to the base of transistor 195.

VOLTAGE CONTROLLED OSCILLATOR AND CONTROL CIRCUITRY THEREFOR, FIG. 5

Reference is now made to FIG. 5 of the drawing wherein there is illustrated a circuit diagram of voltage controlled oscillator 9 and control circuitry therefor. The voltage controlled oscillator basic circuitry is substantially the same as that disclosed in my previouslymentioned copending application, so that a detailed description of the transistors, associated resistors and capacitors is not required herein. The freqquency of oscillator 9 is controlled, inter alia, by the amplitude of the d.c., note indicating voltage derived from the output of sample and hold circuit 8, which is d.c. coupled to oscillator input terminal 201. The frequency of the oscillator is also controlled by the magnitude of: the positive d.c. power supply voltage at terminal 202, a variable voltage at terminal 203, the value of resistor 204 which feeds the voltage at terminal 201 to the oscillator, and the values of substantially equal capacitors 205 and 206 that cross couple the collectors and bases of the oscillator transistors together.

The square wave output frequency of the oscillator, as derived at its output terminal 207, which is connected as an input to frequency divider chain 10, is ex pressed as:

where:

f,, output frequency.

V,, input voltage at terminal 201,

V, DC. power supply voltage at terminal 202,

V voltage at terminal 203,

R value of resistor 204,

C value of each of capacitors 205 and 206. From Equation (1), since V, is greater than V,. the frequency of oscillator 9 is related to variations in the amplitude of the voltage V, in such a manner that as the voltage at terminal 203 increases, the output frequency increases. The voltage at terminal 203 is selectively varied in response to the output of modulation oscillator and to provide flatting effects for certain musical instruments, particularly brasses, which are flatter when first voiced than in a steady state condition. Vibrato modulation of the frequency of oscillator 9 is attained by feeding the output of modulation oscillator 20 to terminal 203. The frequency of modulation oscillator 20 can either be fixed in the range between approximately one to 50 Hertz, normally adjusted for a vibrato rate of approximately seven Hertz, or randomly 16 varied about a mean frequency within this range, In response to the periodic or random variations in the amplitude of the wave derived by modulation oscillator 20, the frequency of oscillator 9 is modulated.

The flatting effect is a transient function of the note indicating signal fed by voltage divider 3 to lead 24. As the pitch of the note increases, the flatting effect is decreased. To these ends, the voltage at terminal 201, indicative of the highest note resulting from depression of key switches 31, is selectively gates to terminal 203 at a time when the tone corresponding with the note is initially being sounded.

For a realistic simulation of the different brass tones, the amount of flatting necessary is different for different instruments. For example, a trombone is initially voiced flatter than a trumpet, whereby it is necessary to have more flatting when simulating a trombone than for simulation of a trumpet. When it is desired to provide the flatting effect for one of the instruments, one of several different positive d.c. voltages is applied to the emitter of PNP transistor 209 by closing one of switches 211 or 221 which are connected to a positive d.c. supply voltage at terminal 212 through resistors 210 and 222 having different values. If switch 211 and resistor 210 are provided for trumpet flatting and switch 221 and resistor 222 for trombone flatting, the value of resistor 222 is smaller than that of resistor 210 to provide a higher emitter current for the trombone and therefore greater trombone flatting.

In response to one of switches 211 or 221 being closed, the difference in voltage amplitude between the note indicating voltage on lead 201 and the emitter voltage of transistor 209 is derived at the collector of transistor 209 which is connected to ground via load resistor 213 that is shunted by the normally cut off emitter collector path of NPN transistor 214. Transistor 214 is driven into a conducting state in response to the positive going trailing edge of the voltage developed at the collector of transistor 49, which is fed to the transistor base via the ac. coupling circuit including capacitor 215 and resistors 216 and 217. A terminal common to the collectors of transistors 209 and 214 is connected to terminal 203 and across relatively small load resistor 208 via relatively large capacitor 218. When transistor 214 is cut off, capacitor 218 is charged to a voltage which is dependent upon the emitter current of transistor 209 and the value of resistor 213 so that as the highest note indicating voltage, V,,, increases the voltage on capacitor 218 decreases.

In response to transistor 214 being transiently activated into a conducting state in response to the positive going, trailing edge transition derived from the collector of transistor 49, capacitor 218 is suddenly discharged through the collector-emitter path of transistor 214. Thereby, the voltage at terminal 203 suddenly decreases by an amount equal to the voltage on capacitor 218 and therefore related to the value of V This applies a negative transient voltage at terminal 203 which causes oscillation to transiently go flat. Capacitor 218 exponentially recharges after transistor 214 returns to a non-cond ucting state. Because the amplitude of the sudden decrease in the voltage at terminal 203 is inversely related to the highest note, greater flatting is provided for lower pitch tones than for higher pitch tones.

BRASS PRESET VOICE FILTERS, FIG. 6

Reference is now made to FIG. 6 of the drawing wherein there is illustrated a circuit diagram of a complete channel 231 for simulation of one brass instrument. the trumpet, as well as control circuitry 233 for the trumpet simulation channel and a further channel 232 for simulating a second brass instrument. e.g., the trombone. Trumpet simulation channel 231 is driven by the 16 output of frequency divider chain 10, while the trombone simulating channel 232 is driven by the 32' output of the frequency divider chain. Channels 231 and 232 are driven in parallel by an output signal derived by envelope shaping network 233 which is responsive to the positive going voltage derived at the collector of transistor 49, FIG. 2. Channels 231 and 232 respond to the signals derived by envelope shapcr 233 to provide simulation of the attack rate, attack tone color change, tone color change as a function of dynamic level, and overall tone quality for the two brass instruments being simulated.

Attack rate and release rate simulation are the same for the two brass instruments, whereby envelope shaping circuit 233 can be utilized to control envelope modulation of both of channels 231 and 232. The attack and release voltage waveform applied by circuit 233 to channels 231 and 232 is illustrated in FIG. 7, wherein the output voltage of circuit 233 is illustrated as a function of time. During the first few. approximately six,

milliseconds after derivation of the positive going. trailing edge at the collector of transistor 49, the voltage developed by circuit 233 increases at a relatively rapid exponential rate, as indicated by line segment 234. After the six millisecond interval has elapsed, the rate of voltage increases of the output of circuit 233 decreases and assumes the exponential relationship indicated by waveform segment 235. Upon release of a note, the waveform derived by circuit 233 decays at a rate indicated by exponential decay wave portion 236.

To enable the wave shape indicated by FIG. 7 to be derived, circuit 233 includes three cascaded NPN transistors 237, 238 and 239 arranged so that the base of each succeeding stage is connected to be driven by the collector of the preceding stage. Transistors 237 and 239 are normally biased to cut off condition, while transistor 238 is normally biased to be conducting.

The base of transistor 237 is dc. coupled via resistor 241 and terminal 242 to the collector of transistor 49 so that in response to a positive voltage being derived at the collector of transistor 49, transistor 237 is driven from its normally cut off state into a conducting state. The resulting decrease in the voltage at the collector of transistor 237 is coupled to the base of transistor 238, causing the latter transistor to be biased into a cut off state. In response to transistor 238 being cut off, capacitor 243, which in combination with diode 242 shunts the collectonemitter path of transistor 238, is charged by the dc power supply voltage connected to terminal 244 via resistor 245 and diode 242. Thereby, the voltage across capacitor 243 increases as indicated by the waveform segment 234.

In response to the voltage across capacitor 243 reaching a predetermined level, the charge rate of the capacitor is decreased since resistor 246 and diode 247 are connected in series from the collector of transistor 238 to a volt dc. power supply at terminal 248. In response to the voltage across capacitor 243 increasing above the voltage drop of diode 247, to a level of 5.5

18 volts, the diode is forward biased so that current from terminal 244 is shunted through resistor 246 and the diode to decrease the charging rate of capacitor 243, as indicated by waveform segment 235. Capacitor 243 continues to charge until the voltage across it reaches a predetermined value, such as 9.5 volts, at which time the capacitor is fully charged.

Capacitor 243 remains charged until all of key switches 31 are deactivated, at which time the voltage at the collector of transistor 49, applied by terminal 242 to the base of transistor 237, drops to a zero level, causing transistor 237 to cut off and transistor 238 to be saturated. Saturation of transistor 238 results in a discharge of capacitor 243 through resistor 249 and the saturated collector-emitter path of transistor 238, to provide waveform segment 236. The voltage variations across capacitor 243 are coupled to the base of transistor 239 and thence to the emitter of the transistor which is a driver for channels 231 and 232.

Waveform segments 234 and 235 control the plural sequentially derived fast and slow attack rates for the instruments simulated by channels 231 and 232, while waveform segment 236 simulates the decay rate of the instruments. It has been found that effective simulation of the instruments can be provided by the attack and decay rates described.

Channels 231 and 232 are substantially the same, with the exception of certain components included in the former which may not necessarily be included in the latter. To provide the different simulation effects, however, the circuits have different component values as required. Because the circuits are substantially the same, channel 231 is described to the exclusion of channel 232; the elements which are not in channel 232, but which are in channel 231, are indicated infra.

When the system is activated to provide simulation of trumpet sounds, the base of NPN transistor 25] is connected to be responsive to the square wave signal derived on the 16' output lead of frequency divider chain 10, as coupled through capacitor 252 and diode 253 which is forward biased in response to the positive voltage applied to its anode by the dc. power supply connected to terminal 254 and resistor 255. If no trumpet simulation is desired, transistor 251 is driven into saturation by connecting the dc. power supply voltage at terminal 212, FIG. 5, through switch 211 and resistor 256 to the transistor base; transistor 251 is prevented from passing a tone signal because it is held in saturation due to base current being supplied from terminal 212 via resistor 256 and switch 211. Switch 211 is the same switch as is illustrated in FIG. 5; it is a single pole, double throw switch arranged so that its contact connects terminal 212 to only one of resistors 210 or 256. Thereby, the trumpet flatting effect is provided only when switch 211 is activated to enable transistor 251 to be responsive to the 16' output of frequency divider chain 10.

The square wave tone signal applied to input capacitor 252 results in a pulse waveform at the collector of transistor 251 which is amplitude modulated in response to the voltage supplied to the base of transistor 239. Envelope modulation of the tone signal occurs because the voltage supplied to the collector of transistor 251 is derived from the emitter of transistor 239, having a waveform as shown by FIG. 7, which enables simulation of the plural attack rates indicated by wave seg ments 234 and 235 and the decay rate indicated by wave segment 236.

The tone signal developed at the collector of transistor 251 is fed via diode 258 to a bandpass filter 259 of the active type. Diode 258 is included to enable the bandpass filter to be decoupled from any voltage variations which might appear at the collector of transistor 251 when trumpet simulation is not performed.

Bandpass filter 259 is of the active type. including NPN transistor 261 and a feedback circuit from the emitter of the transistor to its base, which is respponsive to the signal coupled through diode 258 via capacitor 262. The base of transistor 261 is forward biased by a positive dc. voltage connected to terminal 263 and resistor 264 to a terminal between filter resistor 265 and capacitor 262. The feedback path from the emitter of transistor 261 to its base includes capacitor 266 which provides a first shunt path for the transistor emitter base junction, as well as the series combination of resistor 267 and capacitor 268 which provides a second shunt path for the emitter base junction. The connection between resistor 267 and capacitor 268 is shunted to ground by capacitor 269. The signal derived at the output of bandpass filter 259 is developed across emitter load resistor 270. The values of the components in cluded in bandpass filter 259 are selected so that the filter center frequency is approximately at l2()() Hz.

The filtered output signal of bandpass filter 259 is fed to variable wave shaper 273 which controls tone color during the attack phase of the tone signal applied thereto whereby brightness increases as time increases as a voice is being initially sounded. This effect is achieved by varying the non-linear impedance of circuit 273 so that during the attack phase the high frequencies derived from bandpass filter 259 are attenuated as a variable function of time. To these ends, the

emitter of transistor 261 is connected to resistor 274,

which is connected to ground through a variable nonlinear impedance shunt path including capacitor 275 and diode 276.

When no note is being played, diode 276 is forward biased to provide alow impedance shunt path between resistor 274 and ground, whereby the filter attenuates the high frequencies in the tone signals. Diode 276 is forward biased in response to the relatively high voltage at the collector of transistor 237, which is fed to the anode of diode 276 through diode 277 and a bias control network including series resistors 278 and 279, the junction between which is shunted to ground by capacitor 280. Diode 277 is poled in such a manner that capacitor 280 is charged to 0.75 of the voltage at the collector of transistor 237 in response to transistor 237 being biased to its cut off condition, as exists when no note is depressed and for the first 20 milliseconds after depression of a note.

In response to transistor 237 being forward biased 20 milliseconds after initial depression of a key, diode 277 is back biased and capacitor 280 is discharged through diode 276 and resistor 279. As time progresses, the discharge current decreases and the impedance of diode 276 is increased. until the diode is no longer biased to a conducting state. As the impedance of diode 276 increases, the shunt impedance from resistor 274 to ground increases, reducing the attenuation of the high frequencies in the tone signal passed through circuit 273. For a typical circuit simulating the attack action of a trumpet, the variable wave shaping action provided by diode 276 and capacitor 275 is completed in approximately 200 milliseconds. In response to release of all keys a similar variable filtering effect is provided in 20 the opposite direction in response to diode 276 being forward biased in response to the increasing voltage developed across capacitor 280 from the relatively high voltage at the collector of transistor 237.

It has been found that the tone brightening effect provided by the variable impedance connected between resistor 274 and ground is not as important for trombone simulation as for trumpet simulation. Therefore. in channel 232, the tone brightening effect is not necessarily included and the components associated therewith, i.e., capacitors 275 and 280, resistors 278 and 279, and diodes 276 and 277 can be excluded.

Additional tone color filtering, under control of the musician as a function of tone loudness, is provided by variable wave shaper 282 that receives the tone signal derived from tone color control circuit 273. Dynamic wave shaping by 282 is controlled by the musician depressing expression shoe 18. In response to no depression of expression shoe 18, the high frequencies of the tone signal are passed through circuit 282 with substantial attentuation, while minimum attenuation and dynamic wave shaping of the high frequencies are provided by maximum depression of the expression shoe.

To these ends, circuit 282 includes a variable impedance shunt path across the output of circuit 273. The variable impedance shunt path comprises capacitor 283 and diode 284, having its anode connected to ca pacitor 283 and its cathode grounded. The junction between diode 284 and capacitor 283 is connected to a variable dc. voltage at slider 285 of potentiometer 286 via coupling resistor 287. Slider 285 is controlled by depression of expression shoe 18 so that in response to the expression shoe being completely depressed, the slider picks off a very low or zero dc voltage, but if the expression shoe is not depressed, a maximum d.c. voltage is fed to diode 284 by slider 285. In response to full depression of expression shoe l8, diode 284 is biased to cut off. whereby a high shunt impedance is provided for the high frequencies of the tone signal and maximum brightness is thereby attained. In contrast, diode 284 is forward biased to a great extent if the expression shoe is not depressed at all, to provide a great deal of attenuation for the high frequencies of the tone signal and reduced brightness. The signal developed across capacitor 283 and diode 284 is fed to the output terminal of channel 231 via a coupling network including series resistor 288 and capacitor 289.

The output of channel 231 is combined with the out put signal of channel 232, which provides trombone simulation in response to the 32' tone input, and the attack waveform developed at the emitter of transistor 239, as well as an indication of expression shoe position, as coupled to a potentiometer pick off point within channel 232. The tones derived from channels 231 and 232 are combined in an additive manner to provide complete simulation for the two brass instruments.

FLUTE FILTERS, FIG. 8

Reference is now made to FIG. 8 of the drawing wherein there is illustrated in partially block diagram and partially circuit schematic diagram flute filters 12. There are five separate flute filter channels 201-205, each of which is substantially the same, except for fixed circuit component values associated with different cut off frequencies for the different channels. The five channels 201, 202, 203, 204 and 205 are respectively responsive to the 1 Va, 2 4, 8' and 16 square wave 

1. An audio frequency filter comprising in cascade in the order recited, a first operational amplifier, a two quadrant first multiplier, a second integrating operational amplifier having two inputs, a second two quadrant multiplier, a third integrating operational amplifier, a source of audio signal applied to a first input of said first operational amplifier, means for applying multiplicative voltage factors respectively to both said first multiplier and said second multiplier, means for feeding back output from said second operational amplifier to a second input of said first operational amplifier, means for feeding back output from said third operational amplifier to said first input of said first operational amplifier, and means for deriving separate audio signal outputs from each of said operational amplifiers.
 2. The combination according to claim 1, wherein said first operational amplifier is a unity gain phase inverter.
 3. The combination according to claim 1, wherein the multiplicative voltage factors of said multipliers are the same.
 4. The combination according to claim 1, wherein said multiplicative voltage factors are dc voltages.
 5. The combination according to claim 1, wherein said means for feeding back output from said second and third operational amplifiers to an input of said first operational amplifier include negative feedback loops.
 6. The combination according to claim 5, wherein said operational amplifiers have transfer functions such that said filter has selectively high pass, low pass and band pass characteristics variable as a function of said multiplicative voltages in respect to cut-offs or center frequency.
 7. The combination according to claim 5, wherein each of said operational amplifiers includes a non-phase inverting and a phase inverting input terminal, and wherein the output of each of said amplifiers is proportional to the differences of the voltages applied to said terminals.
 8. The combination according to claim 7, wherein said first operational amplifier includes a positive and a negative input terminal, said first input being a negative input terminal, and wherein said second and third operational amplifiers each includes one grounded terminal.
 9. An audio frequency amplifier, comprising an audio signal input terminal, a first different operational amplifier having resistive feedback and having a first and a second input terminal, means connecting said signal input terminal to said first input terminal, a first multiplier connected in cascade with the output terminal of said first differential operational amplifier, a second operational amplifier having a further input terminal connected to the output of said first multiplier, said second operational amplifier having a capacitive feedback, means deriving a voltage from the output of said second operational amplifier, means applying a variable fraction of said voltage to said second input terminal, a second multiplier connected in cascade with said second operational amplifier, a third operational amplifier connected in cascade with said second multiplier, said third operational amplifier having an output terminal, a negative feedback circuit connected from said last Mentioned output terminal to said first input terminal, means for at will deriving output signal from any one or more of said operational amplifiers, a source of dc control voltage, and means applying said dc control voltage to both said multipliers as multiplicative factors.
 10. The combination according to claim 9, wherein said audio signal input is a tone signal and wherein said source of dc control voltage is a dc voltage representative of the pitch of said tone signal.
 11. The combination according to claim 9, wherein said audio signal input is a tone signal, and wherein said source of dc control voltage is representative of amplitude of said tone signal.
 12. The combination according to claim 9, wherein said audio signal input is a tone signal and wherein said dc control voltage is representative of a tonal characteristic of said tone signal. 